Circuit providing harmonic response rejection for a frequency mixer

ABSTRACT

An apparatus for reducing a harmonic response in an electronic circuit is provided. The apparatus includes an RF input configured to provide a first signal operating at a radio frequency. The apparatus includes a local oscillator configured to produce a second signal operating at a local oscillator (LO) frequency. The apparatus includes a switching mixer configured to mix the first and second signals. The apparatus includes a notch filter comprising an inductor and a capacitor connected in parallel. The notch filter is directly coupled to an input of the switching mixer in series. The notch filter is tuned such that its resonant frequency is a harmonic of the LO frequency signal. In an aspect, the apparatus also includes a transformer configured to provide the first signal. In an aspect the apparatus also includes a second notch filter comprising a second inductor and a second capacitor connected in parallel.

TECHNICAL FIELD

Various exemplary embodiments disclosed herein relate generally toelectronic circuits. In particular, various embodiments relate tofrequency mixers.

BACKGROUND

The proliferation of a vast variety of wireless applications, such ascellular communications, WLAN, GPS, has created co-existence issues.This is especially prevalent in modern, miniaturized cellular handsettransceivers, as such devices can have multiple, concurrently operatingtransmitters and receivers working over a wide frequency range.

There are several mechanisms that can result in spurious response in areceiver. A particularly troublesome one is caused by the harmonicresponse of switching mixers commonly used in transmitter and receiverchains. In a receiver chain, for example, a switching mixerdown-converts an input frequency signal to generate an output signal ata lower frequency, f_(IF). However, while the switching mixerdown-converts the desired signal at f_(LO), the local oscillatorfrequency, it also down-converts harmonic frequencies, to f_(IF),causing contamination of the desired signal.

In view of the foregoing, it would be desirable to provide a circuitthat filters harmonic responses for a switching mixer. In particular, itwould be desirable to produce a low-power, low noise, stable circuitthat rejects harmonics responses in a switching mixer.

SUMMARY

In light of the present need for a harmonic response rejection circuitfor a switching mixer, a brief summary of various exemplary embodimentsis presented. Some simplifications and omissions may be made in thefollowing summary, which is intended to highlight and introduce someaspects of the various exemplary embodiments, but not to limit the scopeof the invention. Detailed descriptions of a preferred exemplaryembodiment adequate to allow those of ordinary skill in the art to makeand use the inventive concepts will follow in the later sections.

In an aspect, an apparatus for reducing a harmonic response in anelectronic circuit is provided. The apparatus includes an RF inputconfigured to provide a first signal operating at a radio frequency. Theapparatus also includes a local oscillator configured to produce asecond signal operating at a local oscillator (LO) frequency. Theapparatus also includes a switching mixer configured to mix the firstand second signals. The apparatus also includes at least one notchfilter that includes an inductor and a capacitor connected to theinductor in parallel. The notch filter is directly coupled to an inputof the switching mixer in series. The notch filter is tuned such thatits resonant frequency is a harmonic of the LO frequency signal.

In an aspect, the apparatus also includes a transformer configured toprovide the first signal. In an aspect the apparatus also includes asecond notch filter comprising a second inductor and a second capacitorconnected to the second inductor in parallel. The second notch filter isdirectly coupled to an input of the switching mixer in series. Thesecond notch filter is tuned such that its resonant frequency is aharmonic of the LO frequency signal. In an aspect, the transformercomprises a first winding and a second winding. Each terminal of thesecond winding is connected to an input of the first and second notchfilters, respectively.

In an aspect, the transformer comprises a double-tuned transformer. Inan aspect, inductors in the first and second notch filters have mutualcoupling.

In an aspect, the apparatus also includes a plurality of N pairs ofnotch filters connected in series. Each of N notch filter pairs is tunedto a separate harmonic of the LO frequency signal. The switching mixeronly receives the radio frequency signal as an output of the N notchfilter pairs.

In an aspect, a method for reducing a harmonic response in an electroniccircuit is provided. The method includes at least one notch filterproviding a first signal operating at a radio frequency. The method alsoincludes a switching mixer receiving a second signal operating at an LOfrequency. The method also includes the switching mixer mixing the firstand second signal.

It should be apparent that, in this manner, various exemplaryembodiments enable an improved switching mixer. Particularly, byproviding an embedded notch filter for a mixer in series, responses of amixer at harmonics of a local oscillator frequency can be rejected witha reduced noise penalty.

BRIEF DESCRIPTION OF THE DRAWINGS

In order to better understand various exemplary embodiments, referenceis made to the accompanying drawings wherein:

FIG. 1 illustrates a wireless device communicating with differentwireless communications systems;

FIG. 2 illustrates an exemplary wireless transceiver;

FIG. 3 illustrates an exemplary stage of a wireless receiver front-end,including a switching mixer;

FIG. 4 illustrates another exemplary stage of a wireless receiverfront-end, including a switching mixer;

FIG. 5 illustrates an exemplary stage of a wireless receiver thatincludes a passive network;

FIG. 6 illustrates an exemplary impedance graph of an exemplary passivenetwork;

FIG. 7 illustrates exemplary responses for various transformerconfigurations; and

FIG. 8 illustrates an exemplary method for producing a signal using theexemplary stage of a wireless receiver.

DETAILED DESCRIPTION

The detailed description set forth below in connection with the appendeddrawings is intended as a description of various exemplary embodimentsof the present invention and is not intended to represent the onlyembodiments in which the present invention may be practiced. Thedetailed description includes specific details for the purpose ofproviding a thorough understanding of various concepts. However, it willbe apparent to those skilled in the art that the present invention maybe practiced without these specific details. In some instances,well-known structures and components are shown in block diagram form inorder to avoid obscuring such concepts. Acronyms and other descriptiveterminology may be used merely for convenience and clarity and are notintended to limit the scope of the invention. The term “exemplary” isused herein to mean “serving as an example, instance, or illustration.”Any design described herein as “exemplary” is not necessarily to beconstrued as preferred or advantageous over other designs.

Several aspects of telecommunication systems will now be presented withreference to various apparatus and methods. These apparatus and methodswill be described in the following detailed description and illustratedin the accompanying drawings by various blocks, modules, components,circuits, steps, processes, algorithms, etc. (collectively referred toas “elements”). These elements may be implemented using electronichardware, computer software, or any combination thereof. Whether suchelements are implemented as hardware or software depends upon theparticular application and design constraints imposed on the overallsystem.

By way of example, an element, or any portion of an element, or anycombination of elements may be implemented with a “processing system”that includes one or more processors. Examples of processors includemicroprocessors, microcontrollers, digital signal processors (DSPs),field programmable gate arrays (FPGAs), programmable logic devices(PLDs), state machines, gated logic, discrete hardware circuits, andother suitable hardware configured to perform the various functionalitydescribed throughout this disclosure. One or more processors in theprocessing system may execute software. Software shall be construedbroadly to mean instructions, instruction sets, code, code segments,program code, programs, subprograms, software modules, applications,software applications, software packages, routines, subroutines,objects, executables, threads of execution, procedures, functions, etc.,whether referred to as software, firmware, middleware, microcode,hardware description language, or otherwise.

Accordingly, in one or more exemplary embodiments, the functionsdescribed may be implemented in hardware, software, firmware, or anycombination thereof. If implemented in software, the functions may bestored on or encoded as one or more instructions or code on acomputer-readable medium. Computer-readable media includes computerstorage media. Storage media may be any available media that can beaccessed by a computer. By way of example, and not limitation, suchcomputer-readable media can comprise random-access memory (RAM),read-only memory (ROM), electronically erasable programmable ROM(EEPROM), compact disk (CD) ROM (CD-ROM), or other optical disk storage,magnetic disk storage or other magnetic storage devices, or any othermedium that can be used to carry or store desired program code in theform of instructions or data structures and that can be accessed by acomputer. Disk and disc, as used herein, includes CD, laser disc,optical disc, digital versatile disc (DVD), and floppy disk where disksusually reproduce data magnetically, while discs reproduce dataoptically with lasers. Combinations of the above should also be includedwithin the scope of computer-readable media.

The word “exemplary” is used herein to mean serving as an example,instance, or illustration. Any embodiment described herein as“exemplary” is not necessarily to be construed as preferred oradvantageous over other embodiments. Likewise, the term “embodiment” ofan apparatus, circuit or method does not require that all embodiments ofthe invention include the described components, structure, features,functionality, processes, advantages, benefits, or modes of operation.

The terms “connected,” “coupled,” or any variant thereof, mean anyconnection or coupling, either direct or indirect, between two or moreelements, and can encompass the presence of one or more intermediateelements between two elements that are “connected” or “coupled”together. The coupling or connection between the elements can bephysical, logical, or a combination thereof. As used herein, twoelements can be considered to be “connected” or “coupled” together bythe use of one or more wires, cables and/or printed electricalconnections, as well as by the use of electromagnetic energy, such aselectromagnetic energy having wavelengths in the radio frequency region,the microwave region and the optical (both visible and invisible)region, as several non-limiting and non-exhaustive examples.

Any reference to an element herein using a designation such as “first,”“second,” and so forth does not generally limit the quantity or order ofthose elements. Rather, these designations are used herein as aconvenient method of distinguishing between two or more elements orinstances of an element. Thus, a reference to first and second elementsdoes not mean that only two elements can be employed, or that the firstelement must precede the second element.

As used herein, the terms “comprises”, “comprising,”, “includes” and/or“including”, when used herein, specify the presence of the statedfeatures, integers, steps, operations, elements, and/or components, butdo not preclude the presence or addition of one or more other features,integers, steps, operations, elements, components, and/or groupsthereof.

Various aspects of circuits for a harmonic rejection circuit for aswitching mixer will now be presented. However, as those skilled in theart will readily appreciate, such aspects may be extended to othercircuit configurations and devices. Accordingly, all references to aspecific application for mixer arrangements, or any component,structure, feature, functionality, or process within a synchronizedwireless device are intended only to illustrate exemplary aspects ofelectronic hardware with the understanding that such aspects may have awide differential of applications.

Various embodiments of hardware with a double-tuned transformer with anotch filter may be used, such as a mobile phone, personal digitalassistant (PDA), desktop computer, laptop computer, palm-sized computer,tablet computer, set-top box, navigation device, work station, gameconsole, media player, or any other suitable device.

FIG. 1 illustrates a wireless device communicating with differentwireless communications systems. FIG. 1 is a diagram 100 illustrating awireless device 110 communicating with different wireless communicationsystems 120, 122. Wireless device 110 can use a switching mixer, forexample, for communications via carrier waves at specified frequenciesvia techniques like phase modulation; other uses of mixers in electronichardware are known to those of skill in the art.

Wireless systems 120, 122 may each be a Code Division Multiple Access(CDMA) system, a Global System for Mobile Communications (GSM) system, aLong Term Evolution (LTE) system, a wireless local area network (WLAN)system, or some other wireless system. A CDMA system may implementWideband CDMA (WCDMA), CDMA 1× or cdma2000, Time Division SynchronousCode Division Multiple Access (TD-SCDMA), or some other version of CDMA.TD-SCDMA is also referred to as Universal Terrestrial Radio Access(UTRA) Time Division Duplex (TDD) 1.28 Mcps Option or Low Chip Rate(LCR). LTE supports both frequency division duplexing (FDD) and timedivision duplexing (TDD). For example, wireless system 120 may be a GSMsystem, and the wireless system 122 may be a WCDMA system. As anotherexample, the wireless system 120 may be an LTE system, and wirelesssystem 122 may be a CDMA system.

For simplicity, diagram 100 shows wireless system 120 including one basestation 130 and one system controller 140, and wireless system 122including one base station 132 and one system controller 142. Ingeneral, each wireless system 120, 122 may include any number of basestations and any set of network entities. Each base station 130, 132 maysupport communication for wireless devices within the coverage of thebase station. Base stations 130, 132 may also be referred to as a NodeB, an evolved Node B (eNB), an access point, a base transceiver station,a radio base station, a radio transceiver, a transceiver function, abasic service set (BSS), an extended service set (ESS), or some othersuitable terminology. Wireless device 110 may also be referred to as auser equipment (UE), a mobile device, a remote device, a wirelessdevice, a wireless communications device, a station, a mobile station, asubscriber station, a mobile subscriber station, a terminal, a mobileterminal, a remote terminal, a wireless terminal, an access terminal, aclient, a mobile client, a mobile unit, a subscriber unit, a wirelessunit, a remote unit, a handset, a user agent, or some other suitableterminology. Wireless device 110 may be a cellular phone, a smartphone,a tablet, a wireless modem, a personal digital assistant (PDA), ahandheld device, a laptop computer, a smartbook, a netbook, a cordlessphone, a wireless local loop (WLL) station, or some other similarfunctioning device.

Wireless device 110 may be capable of communicating with wireless system120 and/or 122. Wireless device 110 may also be capable of receivingsignals from broadcast stations, such as broadcast station 134. Wirelessdevice 110 may also be capable of receiving signals from satellites,such as satellite 150, in one or more global navigation satellitesystems (GNSS). Wireless device 110 may support one or more radiotechnologies for wireless communication such as GSM, WCDMA, cdma2000,LTE, 802.11, etc. The terms “radio technology,” “radio accesstechnology,” “air interface,” and “standard” may be usedinterchangeably.

Wireless device 110 may communicate with a base station in a wirelesssystem via the downlink and the uplink. The downlink (or forward link)refers to the communication link from the base station to the wirelessdevice, and the uplink (or reverse link) refers to the communicationlink from the wireless device to the base station. A wireless system mayutilize TDD and/or FDD. For TDD, the downlink and the uplink may sharethe same frequency, and downlink transmissions and uplink transmissionsmay be sent on the same frequency in different time periods. For FDD,the downlink and the uplink are allocated separate frequencies. Downlinktransmissions may be sent on one frequency, and uplink transmissions maybe sent on another frequency. Some exemplary radio technologiessupporting TDD include GSM, LTE, and TD-SCDMA. Some exemplary radiotechnologies supporting FDD include WCDMA, cdma2000, and LTE.

FIG. 2 is a block diagram 200 of an exemplary wireless device, such aswireless device 110. Wireless device 200 includes a dataprocessor/controller 210, a transceiver 218, and an antenna 290. Thedata processor/controller 210 may be referred to as a processing system.A processing system may include data processor/controller 210 or bothdata processor/controller 210 and memory 216.

Transceiver 218 includes a transmitter (TX) chain 220 and a receiver(RX) chain 250 that support bi-directional communication. TX chain 220and/or RX chain 250 may be implemented with a super-heterodynearchitecture or a direct-conversion architecture. In thesuper-heterodyne architecture, a signal is frequency converted betweenRF and baseband in multiple stages, e.g., from RF to an intermediatefrequency (IF) in one stage, and then from IF to baseband in anotherstage for a receiver. In the direct-conversion architecture, which isalso referred to as a zero-IF architecture, a signal is frequencyconverted between RF and baseband in one stage. The super-heterodyne anddirect-conversion architectures may use different circuit blocks and/orhave different requirements. In the exemplary design shown in FIG. 2,the transmitter chain 220 and the receiver chain 250 are implementedwith the direct-conversion architecture.

In the transmit chain, data processor/controller 210 may process (e.g.,encode and modulate) data to be transmitted and provide the data to adigital-to-analog converter (DAC) 230. DAC 230 converts a digital inputsignal to an analog output signal. The analog output signal is providedto a transmit (TX) baseband (low-pass) filter 232, which may filter theanalog output signal to remove images caused by the priordigital-to-analog conversion by DAC 230. An amplifier (amp) 234 mayamplify the signal from TX baseband filter 232 and provide an amplifiedbaseband signal. An up-converter (mixer) 236 may receive the amplifiedbaseband signal and a TX LO signal from a TX LO signal generator 276.The up-converter 236 may up-convert the amplified baseband signal withthe TX LO signal and may provide an up-converted signal. A filter 238may filter the up-converted signal to remove images caused by thefrequency up-conversion. A power amplifier (PA) 240 may amplify thefiltered RF signal from filter 238 to obtain the desired output powerlevel and provide an output RF signal. The output RF signal may berouted through a duplexer/switchplexer 264.

For FDD, transmitter chain 220 and receiver chain 250 may be coupled toduplexer 264, which may include a TX filter for transmitter chain 220and an RX filter for receiver chain 250. The TX filter may filter theoutput RF signal to pass signal components in a transmit band andattenuate signal components outside the transmit band. For TDD,transmitter chain 220 and receiver chain 250 may be coupled toswitchplexer 264. Switchplexer 264 may pass the output RF signal fromthe transmitter chain 220 to the antenna 290 during uplink timeintervals. For both FDD and TDD, the duplexer/switchplexer 264 mayprovide the output RF signal to the antenna 290 for transmission via awireless channel.

In the receive chain, the antenna 290 may receive signals transmitted bybase stations and/or other transmitter stations and may provide areceived RF signal. The received RF signal may be routed throughduplexer/switchplexer 264. For FDD, the RX filter within the duplexer264 may filter the received RF signal to pass signal components in areceive band and attenuate signal components outside the receive band.For TDD, the switchplexer 264 may pass the received RF signal from theantenna 290 to the receiver chain 250 during downlink time intervals.For both FDD and TDD, the duplexer/switchplexer 264 may provide thereceived RF signal to the receiver chain 250.

Within the receiver chain 250, the received RF signal may be amplifiedby a low noise amplifier (LNA) 252 and filtered by a filter 254 toobtain an input RF signal. A down-converter (mixer) 256 may receive theinput RF signal and an RX LO signal from an RX LO signal generator 286.The down-converter 256 may down-convert the input RF signal with the RXLO signal and provide a down-converted signal. The down-converted signalmay be amplified by an amplifier 258 and further filtered by an RXbaseband (low-pass) filter 260 to obtain an analog input signal. Theanalog input signal is provided to an analog-to-digital converter (ADC)262. The ADC 262 converts an analog input signal to a digital outputsignal. The digital output signal is provided to the dataprocessor/controller 210.

A TX frequency synthesizer 270 may include a TX phase locked loop (PLL)272 and a VCO 274. VCO 274 may generate a TX VCO signal at a desiredfrequency. TX PLL 272 may receive timing information from the dataprocessor/controller 210 and generate a control signal for VCO 274. Thecontrol signal may adjust the frequency and/or the phase of VCO 274 toobtain the desired frequency for the TX VCO signal. TX frequencysynthesizer 270 provides the TX VCO signal to TX LO signal generator276. TX LO signal generator 276 may generate a TX LO signal based on theTX VCO signal received from TX frequency synthesizer 270.

A RX frequency synthesizer 280 may include an RX PLL 282 and a VCO 284.VCO 284 may generate an RX VCO signal at a desired frequency. RX PLL 282may receive timing information from the data processor/controller 210and generate a control signal for VCO 284. The control signal may adjustthe frequency and/or the phase of VCO 284 to obtain the desiredfrequency for the RX VCO signal. RX frequency synthesizer 280 providesthe RX VCO signal to RX LO signal generator 286. The RX LO signalgenerator may generate an RX LO signal based on the RX VCO signalreceived from RX frequency synthesizer 280.

LO signal generators 276, 286 may each include frequency dividers,buffers, etc. LO signal generators 276, 286 may be referred to asfrequency dividers if they divide a frequency provided by TX frequencysynthesizer 270 and RX frequency synthesizer 280, respectively. PLLs272, 282 may each include a phase/frequency detector (PFD), a filter(e.g., a loop filter), a charge pump, a frequency divider, etc. Each VCOsignal and each LO signal may be a periodic signal with a particularfundamental frequency. The TX LO signal and the RX LO signal from LOgenerators 276, 286 may have the same frequency for TDD or differentfrequencies for FDD. The TX VCO signal and the RX VCO signal from VCOs274, 284 may have the same frequency (e.g., for TDD) or differentfrequencies (e.g., for FDD or TDD).

The conditioning of the signals in transmitter chain 220 and receiverchain 250 may be performed by one or more stages of amplifier, filter,up-converter, down-converter, etc. These circuits may be arrangeddifferently from the configuration shown in FIG. 2. Furthermore, othercircuits not shown in FIG. 2 may also be used to condition the signalsin transmitter chain 220 and receiver chain 250. For example, impedancematching circuits may be located at the output of PA 240, at the inputof LNA 252, between antenna 290 and duplexer/switchplexer 264, etc. Somecircuits in FIG. 2 may also be omitted. For example, filter 238 and/orfilter 254 may be omitted. All or a portion of transceiver 218 may beimplemented on one or more analog integrated circuits (ICs), RF ICs(RFICs), mixed-signal ICs, etc. For example, TX baseband filter 232 toPA 240 in transmitter chain 220, LNA 252 to RX baseband filter 260 inreceiver chain 250, PLLs 272, 282, VCOs 274, 284, and LO signalgenerators 276, 286 may be implemented on an RFIC. PA 240 and possiblyother circuits may also be implemented on a separate IC or a circuitmodule.

Data processor/controller 210 may perform various functions for thewireless device. For example, data processor/controller 210 may performprocessing for data being transmitted via transmitter chain 220 andreceived via receiver chain 250. Data processor/controller 210 maycontrol the operation of various circuits within transmitter chain 220and receiver chain 250. Memory 212 and/or memory 216 may store programcodes and data for data processor/controller 210. The memory may beinternal to data processor/controller 210 (e.g., memory 212) or externalto data processor/controller 210 (e.g., memory 216). The memory may bereferred to as a computer-readable medium. An oscillator 214 maygenerate a VCO signal at a particular frequency. A clock generator 215may receive the VCO signal from oscillator 214 and may generate clocksignals for various modules within data processor/controller 210. Dataprocessor/controller 210 may be implemented on one or more applicationspecific integrated circuits (ASICs) and/or other ICs.

FIG. 3 illustrates an exemplary stage of a wireless receiver front-end,including a switching mixer. Circuit 300 includes a low-noise amplifiercomprising an LC matching circuit 303, an LC trapping circuit 305 andamplifying devices 307 a-307 b, transformer 309, notch filter 311,harmonic reject mixer 313 and trans-impedance amplifier (TIA) 315.Circuit 300 can be implemented, for example, as multiple components ofRX chain 250 for wireless receiver 200. In some embodiments, circuit 300is configured to be implemented as multiple components in RX chain(e.g., LNA 252, filter 254, and mixer 256). The illustrative embodimentof circuit 300 includes multiple rejection features that can be usedindependently or in tandem to reject undesired signals at one or moreharmonics of the local oscillator frequency, such as LO frequency signalproduced by RX LO signal generator 286.

The Low-Noise Amplifier (LNA) can comprise, for example, LC matchingcircuit 303, LC trapping circuit 305 and amplifying devices 307 a-307 b.In some embodiments, the LNA can comprise other elements of the circuit.In the illustrative embodiment, for example, the LNA can be a cascodeamplifier that provides an output current to transformer 309. Thecascode configuration can, for example, provide improved isolationbetween the input and output port of the LNA and can improve, forexample, gain and stability for the circuit.

LC matching circuit 303 can include an L-section of an impedancematching circuit, which can include a shunt-connected capacitor 303 band a series-connected inductor 303 a. The output of inductor 303 a canbe connected to the input of amplifying devices 307 a-307 b. Duringoperation, capacitor 303 b provides a low impedance to ground; theimpedance to ground decreases as the frequency of the signal increases.Conversely, inductor 303 a has an increased impedance as the frequencyincreases. The inductor-capacitor combination 303 a-303 b can suppressspecific undesired high-frequency signals, such as signals withfrequencies falling at harmonics of the LO frequency. In someembodiments, multiple LC matching sections can be placed in a cascade;this can increase rejection of high frequencies at this stage of thecircuit.

In some embodiments, traps, such as a parallel and/or series traps, canbe used to reject one or more specified frequencies. A person ofordinary skill would be aware of inclusion of such traps before theinput of amplifying devices 307 a-307 b. The addition of suppressioncomponents, such as LC matching circuit 303, or combinations of matchingcircuits and/or traps, can be weighed against factors like cost, printedcircuit board (PCB) area, and noise performance degradation that isassociated with use of the suppression components.

LC trapping circuit 305 can include a source degeneration inductor (Ls)305 a and a source degeneration capacitor (C_(S)) 305 b. In someembodiments, only source degeneration inductor 305 a is included in thecircuit. Source degeneration capacitor 305 b can be added to resonate ata harmonic of the local oscillator frequency and suppress the gain ofthe LNA at that specific LO harmonic frequency. The addition of sourcedegeneration capacitor 305 b can be weighed against overall stability ofthe LNA.

Amplifying devices 307 a-307 b can be transistors, such asmetal-oxide-semiconductor field-effect transistors (MOSFETs) or othercomponents that amplify a signal received at the input of 307 a bydrawing an output current from supply voltage V_(dd) through the primarywinding of transformer 309. The configuration of amplifying devices 307a-307 b can be arranged, for example, to reduce Miller capacitance andincrease the bandwidth of the LNA.

Transformer 309 can be a transformer that outputs a current in itssecondary winding that is based on the current at its primary winding.In the illustrative embodiment, for example, transformer 309 is asingle-tuned transformer that includes a tunable capacitor at itsprimary winding. In some embodiments, the capacitor can be anothercomponent that provides an adjustable capacitance, such as a varactor.In some embodiments, the secondary winding is connected in parallel withnotch filter 311.

Notch filter 311 can, for example, act as a trap resonating at aharmonic of the local oscillator frequency to shunt the secondarywinding of transformer 309. Notch filter 311 can provide rejection atthe configured resonant frequency. In some embodiments, notch filter 311can be added to provide a resonating trap at a designated harmonic. Theamount of rejection provided by notch filter 311 can be dependent on theratio between the input impedance of mixer 313 and the trap impedance atthe designated harmonic frequency. In some embodiments, such as wheremixer 313 is a passive switching mixer, for example, the noisecontributed by the trans-impedance operational amplifier (V_(ntia) 315b) to the output can generally be proportional to the rejection providedby notch filter 311.

Mixer 313 can mix the amplified radio frequency signal received from theoutput of the secondary winding of transformer 309 and a localoscillator frequency signal to produce an output signal. In someembodiments, mixer 313 produces a baseband signal current (i_(bb)) asthe output signal. In some embodiments, mixer 313 can comprise an IQmixer that produces separate in-phase (I) and quadrature-phase (Q)signals. In some embodiments, mixer 313 can comprise a harmonicrejection mixer that suppresses the harmonic response that would beproduced when employing a switching mixer. In such embodiments, harmonicrejection mixer 313 can receive a multi-phase local oscillator signal inlieu of a single-phase LO signal. Generation of a multi-phase localoscillator signal can require more physical space on a circuit chip andcan consume more current than the generation of a correspondingsingle-phase local oscillator signal. Use of the multi-phase signal canbe weighed against factors like cost and power consumption. In someembodiments, harmonic rejection mixer 313 can comprise an active mixer.

Trans-impedance operational amplifier (TIA) 315 includes an operationalamplifier 315 a, resistors R_(f) and capacitors C_(f). TIA 315 canconvert an input baseband current received from mixer 313 to produce anoutput voltage based on the input current. In the illustrativeembodiment, for example, TIA 315 can convert the input baseband currenti_(bb) into baseband voltage V_(bb). During operation, operationalamplifier 315 a can contribute noise to the output baseband voltageV_(bb). The noise generated by operational amplifier 315 a can berepresented by input noise voltage source V_(ntia) 315 b. V_(ntia) 315 bcan be amplified and appear at TIA 315 output (V_(no) _(_) _(tia))according to a ratio between the impedance of the feedback componentsR_(f) and C_(f) (Z_(f)=1/(1/Z_(Rf)+1/Z_(Cf))) and the baseband impedanceZ_(bb) looking back to the output of mixer 313. The impedance offeedback components comprising of R_(f) and C_(f) can be characterizedas:

$Z_{f} = \frac{1}{\frac{1}{Z_{rf}} + \frac{1}{Z_{cf}}}$

In the illustrative embodiment, where mixer 313 is a mixer, the basebandimpedance Z_(bb) is proportional to RF impedance Z_(rf) presented tomixer 313 input at LO fundamental and harmonic frequencies. Thus, if:

$V_{no\_ tia}^{2} \propto {{1 + \frac{Z_{f}}{Z_{bb}}}}^{2}$

The TIA noise contribution is proportional to the reduced harmonicresponse if notch filter 311 is used in combination with mixer 313. Theharmonic response rejection has to be chosen such that TIA noisecontribution to baseband output does not grow too large.

Depending on potential tradeoffs related to issues like physical areaand noise, circuit 300 can include one or more components, such as LCmatching circuit 303, LC trapping circuit 305, notch filter 311, and/orharmonic reject mixer 313 to reject undesired interference that areharmonics of the LO frequency.

FIG. 4 illustrates another exemplary stage of a wireless receivingfront-end, including a switching mixer 413. Circuit 400 can receive aninput signal from a LNA, such as the LNA in circuit 300, and produce anoutput baseband voltage (V_(bb)). Circuit 400 includes a transformer409, notch filters 411-412, switching mixer 413, and TIA 415. Circuit400 is similar to circuit 300 and can be configured to be implemented asmultiple components included in RX chain 250.

Transformer 409 can include a primary winding and a secondary winding.In some embodiments, transformer 409 can be a single-tuned ordouble-tuned transformer. When transformer 409 is tuned, it includes atunable passive element, such as a tunable capacitor or varactor. Theinductive reactance of transformer 409 can be tuned out, with thetunable capacitor providing a shunt capacitance at the resonantfrequency. In the illustrative embodiment, for example, transformer 409is a double-tuned transformer that includes a first tunable capacitor(C_(P)) 409 a and a second tunable capacitor (C_(S)) 409 b.

Transformer 409 can have a coupling co-efficient (k) between the primaryand secondary windings. In the illustrative embodiment, for example,transformer 409 can have a relatively loose coupling co-efficient (e.g.,0.6≦k≦0.65). The loose coupling can allow transformer 409 to bedouble-tuned at the primary winding by C_(P) 409 a and the secondarywinding by C_(s) 409 b, respectively. The double tuning can produce animproved harmonic response rejection by giving better selectivity toreject out-of-band harmonics, as well as greater control over thein-band frequency response and output impedance of the LNA of FIG. 3(Z_(rf)).

In some embodiments, transformer 409 can produce a differential signal,with the secondary winding sending the signals to notch filters 411 and412, respectively. Notch filters 411-412 can be tunable LC traps (i.e.,tank circuits) that are tuned to have their resonant frequency equalthat of a harmonic of the LO frequency. Traps 411-412 can be placedbetween transformer 409 and switching mixer 413 to suppress inputs toswitching mixer 413 that fall at a harmonic of the LO frequency. Trap411 can include a tunable capacitor 411 a and inductor 411 b, while trap412 can include tunable capacitor 412 a and inductor 412 b. Traps411-412 can be in series and can each receive a component of thedifferential signal produced by the secondary winding of transformer409. In some embodiments, each of traps 411-412 can be tuned to havetheir resonant frequency fall at a harmonic of the LO frequency. In suchinstances, traps 411-412 may each be tuned to the same harmonicfrequency. In some embodiments, tunable capacitors 411 a, 412 a can betuned to resonate at a different harmonic frequency. In someembodiments, capacitors 411 a, 412 a can be tuned based on changes tothe LO frequency. In some embodiments, inductors 411 b, 412 b can havemutual coupling. The mutual coupling between inductors 411 b, 412 b canbroaden the notch bandwidth and can therefore increase the Q factor andreduce physical area associated with traps 411-412.

In the illustrative embodiment, for example, traps 411-412 can each betuned to have their resonant frequency fall at the third harmonic of theLO frequency (3*f_(LO)). As traps 411-412 are tuned to have theirresonant frequencies at the third harmonic, each trap 411-412 provideshigh impedance for input at the third harmonic frequency as seen byswitching mixer 413 such that the input current (i_(rf)) is suppressedat the input of switching mixer 413 at the third harmonic frequency. Aswill be discussed in further detail in relation to FIG. 5, circuit 400can include a passive network comprising a plurality of notch filterpairs, where each pair is tuned such that each tank pair 411-412independently suppresses different harmonic frequencies.

Switching mixer 413 can be a mixer that produces a current based on aninput current received from the transformer 409. The output currentproduced by switching mixer 413 can be based on an input radio frequencysignal (i_(rf)) and an input local oscillator frequency (f_(LO)).Switching mixer 413 can be implemented as an IQ mixer that producesseparate I (in-phase) and Q (quadrature-phase) signals. In someembodiments, the LO frequency can have a specified duty cycle, such as a25% duty cycle. In some embodiments, switching mixer 413 can receive asecond input current that is based on the local oscillator frequency(f_(LO)) and produces the output signal based on the input radiofrequency current and the input local oscillator current.

In some embodiments, switching mixer 413 can be a passive mixer. In someembodiments, switching mixer 413 can be a balanced mixer, such as, forexample, a single-balanced mixer or a double-balanced mixer. Thebalanced mixer can be configured such that even harmonic responses aregreatly reduced. In the illustrative embodiment, for example, switchingmixer 413 can be a double-balanced mixer that receives a differentialinput signal from transformer 409 and produces a differential outputsignal for TIA 415. In some embodiments, switching mixer 413 can be asingle-balanced mixer that receives the single-ended input signal fromtransformer 409 or the LNA directly and produces a differential outputsignal for TIA 415.

Switching mixer 413 can produce a baseband signal current (i_(bb)) thatis based on a combination of the radio frequency current and the localoscillator signal. For example, when used as part of an RX chain 250,switching mixer 413 can down-convert input signals to produce an outputbaseband current that has an output frequency equal to that of thedifference between the input radio frequency and the LO frequency (i.e.,f_(bb)=|f_(rf)−f_(LO)|).

When the switching mixer 413 down-converts the input signal, frequenciesat harmonics of the LO frequency at the input can also be folded intothe output signal (this process is conventionally known as “noisefolding”). In some embodiments, the configuration of traps 411-412 tosuppress harmonic currents can lower the effects of noise folding, asthe magnitude of harmonic currents folded into the output current arelowered. In some embodiments, additional traps 411-412 are included incircuit 400 to suppress frequencies falling at other harmonics; this canreduce the noise folding effects for the other harmonic frequencies.

Trans-impedance amplifier (TIA) 415 can include an operational amplifier415 a and tuning component pairs R_(f) and C_(f). TIA 415 is similar toTIA 315 and can convert an input baseband signal current to produce anoutput signal voltage. In some embodiments, the output signal voltagecomprises a differential signal.

In some instances op-amp 415 a can contribute noise to the outputvoltage V_(bb). The noise generated by operational amplifier 415 a canbe represented by voltage source V_(ntia) 415 b. As discussed above inrelation to V_(ntia) 315 b, the magnitude of the noise is proportionalto the ratio of the feedback impedance of TIA 415 and the basebandimpedance (Z_(f)/Z_(bb)). When switching mixer 413 comprises a passivemixer, the baseband current i_(bb) (and the associated basebandimpedance Z_(bb)) is a result of the down-conversion of all RF currentsat harmonics of the LO frequency (f_(LO), 2*f_(LO), 3*f_(LO), etc.) byswitching mixer 413. The implementation of traps 411-412 can increasethe total baseband impedance, as the impedance Z_(rf) at the specifiedharmonic frequency is increased due to resonance of tanks 411-412 atthat frequency. In the illustrative embodiment for example, themagnitude of baseband impedance increases when traps 411-412 are tunedto resonate at the third harmonic of the LO frequency. The increase intotal baseband impedance reduces the TIA output noise V_(no) _(_) _(tia)contributed by V_(ntia) 415 b.

In some embodiments, circuit 400 can include a loosely-coupled,double-tuned transformer 409 in conjunction with series-connected,series harmonic-resonant traps 411-412, which can improve rejection oflocal oscillator harmonics frequencies. In some embodiments, theharmonic rejection is greater than that of the series notch filter 311of circuit 300.

FIG. 5 illustrates an exemplary stage of a wireless receiver thatincludes a passive network for harmonic rejection. Circuit 500 issimilar to circuit 400 and includes transformer 509, passive network 511(including traps 511 a-511 f), and switching mixer 513.

As discussed above in relation to traps 411-412 in FIG. 4, circuit 500can include passive network 511 placed between the output of transformer509 and the input of switching mixer 513 to suppress one or morefrequencies that fall at harmonics of the LO frequency. In someembodiments, switching mixer 513 can be balanced and can suppressfrequencies falling at even harmonics of the LO frequency (i.e.,2*N*f_(LO), where N=1, 2, 3, etc.).

In such instances, passive network 511 can include one or more seriespairs of harmonic frequency traps 511 a-511 f, with each pair tuned tosuppress frequencies at odd harmonics of the LO frequency (e.g.,3*f_(LO), 5*f_(LO), 7*f_(LO), etc.). Each pair of series traps 511 a-511f can be tuned such that their resonant frequency is a frequency thatfalls at a harmonic of the LO frequency. When connected in series, thecascade configuration of traps can allow each trap pair 511 a-511 f tosuppress a specified frequency while allowing other frequencies to passthrough without significant suppression. Passive network 511 can beconfigured to suppress multiple harmonic frequencies to improve harmonicfrequency rejection and reduce the noise component of an output signalfrom mixer 513 and a TIA after it.

FIG. 6 illustrates an exemplary impedance graph for an exemplary passivenetwork 511. Impedance graph 600 can correspond to the relativeimpedance seen by the radio frequency current i_(rf) at specificfrequencies when traversing through passive network 511.

Graph 600 can correspond to a configuration of circuit 400, 500 thatincludes a cascade of series harmonic frequency trap pairs 411-412, 511a-511 f that are tuned to have resonant frequencies falling at multipleharmonics of the LO frequency. In some embodiments, multiple seriesharmonic trap pairs 511 a-511 f can be included in passive network 511,one trap pair for each of N harmonics above the fundamental frequency.In some embodiments, mixer 413, 513 can be a balanced mixer andsuppresses even harmonic responses; in such instances, passive network511 can include only series harmonic trap pairs 511 a-511 f for each oddharmonic frequency above the fundamental frequency.

As seen by peaks 603-609, the magnitude of the RF impedance (|Z_(rf)|)is greatest at frequencies falling at the harmonics of the LO frequency.In the illustrative embodiment, for example, mixer 413 can be a passivemixer; the increase of |Z_(rf)| can increase the output basebandfrequency Z_(bb) and can reduce the noise component of an outputbaseband voltage V_(bb) produced by the TIA 415.

FIG. 7 illustrates exemplary responses for various configurations. Graph700 illustrates S-Parameter frequency responses (here, S₂₁) fordifferent configurations of circuit 400 including an LNA. Response 701illustrates a response for a configuration including a double-tunedtransformer 409 and trap pairs 411-412. Response 703 illustrates aresponse for a configuration including a double-tuned transformer 409,but excluding trap pairs 411-412. Response 705 includes a response for aconfiguration including a single-tuned transformer 409 that has a highcoupling coefficient (k=0.85) while excluding traps 411-412.

As discussed in relation to transformer 409, the response of the tankcircuit formed by the primary and secondary winding of transformer 409with the specified capacitance will be at least partially based on thecoupling co-efficient of the transformer. A strong coupling (i.e., highcoupling co-efficient: 0.8≦k≦0.95) can push one of the resonances to ahigh frequency, broadening the bandwidth. In contrast, a weaker coupling(i.e., lower coupling co-efficient: 0.6≦k≦0.7) can result in betterharmonic frequency rejection.

Graph 700 illustrates the S₂₁ frequency responses for specificconfigurations of transformer 409 and trap pairs 411-412. As shown byresponses 701-705, while the S-Parameter response is virtually identicalat the first harmonic frequency (f_(LO)=1.829 GHz), the amount ofattenuation greatly differs at the third harmonic frequency. While asingle-tuned transformer (response 705) only results in 20 dB ofattenuation (35.5 dB of rejection), a configuration that includes adouble-tuned transformer provides 24.8 dB (40.3 dB of rejection) ofattenuation (response 703) and a configuration that includes adouble-tuned transformer and a harmonic trap pair for 3*f_(LO) providesover 38 dB of attenuation (53.5 dB of rejection).

FIG. 8 illustrates an exemplary method for producing a signal using theexemplary mixing stage of a wireless receiver. Method 800 can beperformed, for example, by circuit 400, 500 to provide a basebandvoltage as part of a receiver (RX) chain 250 in wireless receiver 200while attempting to reject input interference at LO harmonicfrequencies, prevent noise folding and reduce TIA output noise.

Method 800 can start at 801 and proceed to step 803, where the tankcircuit formed by transformer 409 are tuned. For example, tunablecapacitors C_(p) 409 a and C_(s) 409 b of transformer 409 can be tunedsuch that the primary and secondary windings of transformer 409 resonateat the same frequency.

In step 805, harmonic trap circuits 411-412 can be tuned. In someembodiments, each trap 411, 412 in a trap circuit pair 411-412 isconnected in series between an output of transformer 409 and an input ofmixer 413, with each trap circuit 411, 412 being tuned to have the sameresonant frequency. The resonant frequency can be tuned in step 805 tobe a frequency that falls at a harmonic of the LO frequency.

In some embodiments, the mixing circuit can include a passive network511 comprising a cascade configuration of series harmonic trap pairs411-412 that are connected in series. In such instances, each of thetrap pairs 411-412 can be tuned independently to have a resonantfrequency that falls at a different harmonic of the LO frequency. Insome embodiments, passive network 511 can include trap pairs 411-412whose resonant frequencies are tuned to only odd harmonics of the LOfrequency. This can be done, for example, when the mixing circuitincludes a balanced mixer 413, 513 that rejects even harmonics of the LOfrequency.

Once components of the circuit are tuned, circuit 400 can receive aninput signal at step 807. In some embodiments, the input signal is an RFsignal received from an amplifier, such as the LNA in circuit 300 or LNA252 of RX chain 250. In some embodiments, the input signal can be acarrier wave received by antenna 290. In some embodiments, transformer409 can receive the input signal as a differential signal. In step 809,transformer 409 produces an RF signal for mixer 413. In someembodiments, the secondary winding of transformer 409 produces adifferential signal.

Once transformer 409 produces the RF signal, the RF signal in step 811is sent as a current through passive trap pairs 411-412. In someembodiments, when the RF signal is a differential signal, each componentsignal that comprises the differential signal is sent through one of thetraps 411, 412 comprising trap pair 411-412. In some embodiments, the RFcurrent is sent through passive network 511, with each component signalpassing through multiple trap pairs 411-412 connected in series in acascade configuration. Trap pair 411-412 is configured to suppresscurrents at their tuned resonant frequency. In such instances, thecurrent sent through the passive traps are attenuated at the specifiedresonant frequency.

In step 813, mixer 413 produces an intermediate frequency (IF) signalbased on the RF current signal received from transformer 409 and aninput LO signal at the LO frequency. In some embodiments, mixer 413 is abalanced mixer that produces the baseband output as a differentialsignal. In some embodiments, mixer 413 is a double-balanced mixer thatreceives the differential RF current (i_(rf)) and produces adifferential output baseband current signal (i_(bb)). In someembodiments, the output is sent to a trans-impedance amplifier (TIA) 415that produces an output baseband voltage signal (V_(bb)) based on thebaseband current signal. Once the output signal is produced, method 800can end at step 815.

It is understood that the specific order or hierarchy of steps in theprocesses/flow charts disclosed is an illustration of exemplaryapproaches. Based upon design preferences, it is understood that thespecific order or hierarchy of steps in the processes/flow charts may berearranged. Further, some steps may be combined or omitted. Theaccompanying method claims present elements of the various steps in asample order, and are not meant to be limited to the specific order orhierarchy presented.

The previous description is provided to enable any person skilled in theart to practice the various aspects described herein. Variousmodifications to these aspects will be readily apparent to those skilledin the art, and the generic principles defined herein may be applied toother aspects. Thus, the claims are not intended to be limited to theaspects shown herein, but is to be accorded the full scope consistentwith the language claims, wherein reference to an element in thesingular is not intended to mean “one and only one” unless specificallyso stated, but rather “one or more.” The word “exemplary” is used hereinto mean “serving as an example, instance, or illustration.” Any aspectdescribed herein as “exemplary” is not necessarily to be construed aspreferred or advantageous over other aspects.” Unless specificallystated otherwise, the term “some” refers to one or more. Combinationssuch as “at least one of A, B, or C,” “at least one of A, B, and C,” and“A, B, C, or any combination thereof” include any combination of A, B,and/or C, and may include multiples of A, multiples of B, or multiplesof C. Specifically, combinations such as “at least one of A, B, or C,”“at least one of A, B, and C,” and “A, B, C, or any combination thereof”may be A only, B only, C only, A and B, A and C, B and C, or A and B andC, where any such combinations may contain one or more member or membersof A, B, or C. All structural and functional equivalents to the elementsof the various aspects described throughout this disclosure that areknown or later come to be known to those of ordinary skill in the artare expressly incorporated herein by reference and are intended to beencompassed by the claims. Moreover, nothing disclosed herein isintended to be dedicated to the public regardless of whether suchdisclosure is explicitly recited in the claims. No claim element is tobe construed as a means plus function unless the element is expresslyrecited using the phrase “means for.”

What is claimed is:
 1. An apparatus for reducing a harmonic response inan electronic circuit, the apparatus comprising: an RF input configuredto provide a first signal operating at a radio frequency; a localoscillator configured to produce a second signal operating at a localoscillator (LO) frequency; a switching mixer configured to mix the firstand second signals; and a first notch filter in a path between the RFinput and the switching mixer comprising: an inductor, and a capacitorconnected to the inductor, the first notch filter being directly coupledto an input of the switching mixer, wherein the first notch filter istuned such that its resonant frequency is a harmonic of the LO frequencysignal.
 2. The apparatus of claim 1, further comprising: a transformerconfigured to receive the first signal from the RF input.
 3. Theapparatus of claim 2, further comprising: a second notch filter,comprising: a second inductor, and a second capacitor connected to thesecond inductor in parallel, the second notch filter being directlycoupled to an input of the switching mixer in series, wherein the secondnotch filter is tuned such that its resonant frequency is a harmonic ofthe LO frequency signal, wherein the transformer comprises: a firstwinding, and a second winding, wherein each terminal of the secondwinding is connected to an input of the first and second notch filters,respectively.
 4. The apparatus of claim 3, wherein each capacitor in thefirst and second notch filters are tunable to configure the first andsecond notch filters to have the resonant frequency of each notch filterfall at a harmonic of the LO frequency signal.
 5. The apparatus of claim3, wherein the first signal comprises a differential signal.
 6. Theapparatus of claim 3, rein inductors in the first and second notchfilters have mutual coupling.
 7. The apparatus of claim 2, wherein thetransformer comprises a double-tuned transformer.
 8. The apparatus ofclaim 7, wherein the transformer is configured to have a couplingco-efficient below 0.7.
 9. The apparatus of claim 1, further comprising:a trans-impedance amplifier directly coupled to the switching mixerconfigured to produce a baseband voltage from a baseband currentproduced by the switching mixer.
 10. The apparatus of claim 1, whereinthe first notch filter is tuned to have a resonant frequency fall at athird harmonic of the LO frequency signal.
 11. The apparatus of claim 1,wherein the LO frequency signal comprises a signal with a 25% dutycycle.
 12. The apparatus of claim 1, further comprising: at least onepair of notch filters connected in series with the first notch filler,the at least one pair of notch filters and the first notch filter beingpart of a plurality of N pairs of notch filters connected in series,wherein each of N notch filter pairs is tuned to have a resonantfrequency fall at a separate harmonic of the LO frequency signal,wherein the switching mixer receives a radio frequency signal as anoutput of the N notch filter pairs.
 13. The apparatus of claim 1,further comprising: an antenna configured to: receive a carrier signal,and provide, to the RF input, the first signal comprising the carriersignal.
 14. The apparatus of claim 1, wherein the switching mixerproduces a down-converted signal.
 15. The apparatus of claim 1, whereinthe switching mixer comprises a passive mixer.
 16. A method for reducinga harmonic response in an electronic circuit, the method comprising:providing, by a first notch filter, a first signal operating at a radiofrequency, the first notch filter comprising an inductor and a capacitorconnected to the inductor and wherein the first notch filter is in apath between an RF input and a switching mixer; receiving, by theswitching mixer, a second signal operating at a local oscillator (LO)frequency, and mixing, by the switching mixer, the first and secondsignals, wherein the first notch filter is directly coupled to an inputof the switching mixer and the first notch filter is tuned such that itsresonant frequency is a harmonic of the LO frequency.
 17. The method ofclaim 16, further comprising: receiving, by a transformer, the RE input;and producing, by the transformer, a transformer output signal, whereinthe first notch filter modifies the transformer output signal.
 18. Themethod of claim 17, wherein the transformer comprises a double-tunedtransformer.
 19. The method of claim 18, further comprising: configuringthe transformer to have a coupling co-efficient below 0.7.
 20. Themethod of claim 17, wherein the switching mixer comprises a passivemixer.
 21. The method of claim 17, further comprising: providing, by asecond notch filter, the first signal, wherein the second notch filtercomprises a second inductor and a second capacitor connected to thesecond inductor in parallel and is directly coupled to an input of theswitching mixer in series and the second notch filter is tuned such thatits resonant frequency is a harmonic of the LO frequency signal, furtherwherein the transformer comprises a first winding and a second winding,each terminal of the second winding being connected to an input of thefirst and second notch filters, respectively.
 22. The method claim 21,further comprising: tuning each capacitor in the first and second notchfilters to configure the first and second notch filters to have theresonant frequency of each notch filter fall at a harmonic of the LOfrequency signal.
 23. The method of claim 21, wherein the first signalcomprises a differential signal.
 24. The method of claim 16, furthercomprising: receiving, by a trans-impedance amplifier directly coupledto the switching mixer, a baseband current based on the mixed first andsecond signals; and converting, by the trans-impedance amplifier, thebaseband current into a baseband voltage.
 25. The method of claim 16,further comprising: tuning the first notch filter to have a resonantfrequency fall at a third harmonic of the LO frequency signal.
 26. Themet d of claim 21, wherein inductors in the first and second notchfilters have mutual coupling.
 27. The method of claim 16, wherein the LOfrequency signal comprises a signal with a 25% duty cycle.
 28. Themethod of claim 16, further comprising: tuning a plurality of N pairs ofnotch filters connected in series, each of N notch filter pairs beingtuned to have a resonant frequency fall at a separate harmonic of the LOfrequency signal; and receiving, by the switching mixer, the radiofrequency signal as an output of the N notch filter pairs.
 29. Themethod of claim 16, further comprising: receiving, by an antenna, acarrier signal, and providing, by the antenna to the RF input, the firstsignal comprising the carrier signal.
 30. The method of claim 16,further comprising: producing, by the switching mixer, a down-convertedsignal based on the mixed first and second signals.
 31. An apparatus forreducing a harmonic response in an electronic circuit, the apparatuscomprising: means for providing, by at least one notch filter, a firstsignal operating at a radio frequency, the at least one notch filtercomprising an inductor and a capacitor connected to the inductor andwherein the at least one notch filter is in a path between an RF inputand a switching mixer; means for receiving, by the switching mixer, asecond signal operating at a local oscillator (LO) frequency, and meansfor mixing, by the switching mixer, the first and second signals,wherein the at least one notch filter is directly coupled to an input ofthe switching mixer and the at least one notch filter is tuned such thatits resonant frequency is a harmonic of the LO frequency.
 32. Anapparatus for reducing a harmonic response in an electronic circuit, theapparatus comprising: means for configuring an RF input to provide afirst signal operating at a radio frequency; a local oscillatorconfigured to produce a second signal operating at a local oscillator(LO) frequency; means for configuring a switching mixer to mix the firstand second signals; and means for implementing at least one notch filtercomprising: an inductor, and a capacitor connected to the inductor inparallel, the at least one notch filter being directly coupled to aninput of the switching mixer in series, wherein the notch filter istuned such that its resonant frequency is a harmonic of the LO frequencysignal.